The existing architectures of telecommunications systems are designed to be adaptable to a number of communication standards. They are referred to as multistandard architectures. This is an essential property for enabling a system to operate in multiple geographical regions, using multiple networks. To provide this function, existing systems are actually based on a set of subsystems connected in parallel, and switched according to the scenario for their use.
In particular, the transceiver of a multistandard system uses as many radio frequency filters or duplexers as are required by the number of standards to be met. The filters and duplexers have the property of selecting a fraction of the frequency spectrum, namely the fraction considered to be “useful” to the system because it contains the information to be processed.
At the present time, these filters are mostly produced by means of piezoelectric technology, and are classed in two major groups, namely filters using surface acoustic wave (“SAW”) resonators and filters using bulk acoustic wave (“BAW”) resonators. Filters of this type are specified in telecommunications systems because they are currently relatively easy to produce by integration methods, and therefore occupy a minimal space (a few square millimeters) at an economically acceptable cost.
However, these components operate by using a material and an architecture which require a central frequency and a bandwidth for each one.
With the multiplication of telecommunications standards and distinctive geographical features, the number of these filtering components in on-board systems is tending to increase. It will now be very useful to find a way of making filters variable in respect of their central frequency and their bandwidth, with the aim of reducing the number of filters, and ultimately using only a single filter.
Usually based on a material having piezoelectric properties, SAW/BAW filters make use of the electromechanical conversion of the energy contained in acoustic resonators in order to provide a filtering function.
Whereas the SAW filter uses acoustic vibrations confined to the surface of a piezoelectric substrate, the BAW resonator operates by the vibration within its thickness of a thin layer of piezoelectric material sandwiched between two electrodes.
FIGS. 1a, 1b and 1c show the behavior of a BAW resonator.
More precisely, FIG. 1a shows a conventional stack of materials for the production of a BAW resonator of the “solidly mounted resonator” type, comprising, on the surface of a substrate S, a stack of layers Ci, acting as a reflector, and a layer of piezoelectric material Mpiezo between two electrodes Ei and Es, covered with a dielectric layer I. FIG. 1b shows the first-order equivalent circuit of a piezoelectric resonator and FIG. 1c shows the resonance and antiresonance frequencies of the piezoelectric resonator.
In both the SAW and the BAW types, this acoustic phenomenon is interpreted in electrical terms as the arrangement of an RLC resonator in parallel with a capacitor. This model, very widely discussed in the literature, is commonly designated by the abbreviation “BVD”, for “Butterworth-Van Dyke”. It then has a series resonance (quasi-short-circuit) and an antiresonance (quasi open-circuit). The frequency distance separating the resonance from the antiresonance is characterized by the electromechanical coupling coefficient of the resonator, defined by the following equation:
      k    eff    2    =            π      2        ⁢                  f        s                    f        p              ⁢          tan      ⁡              (                              π            2                    ⁢                                                    f                p                            -                              f                s                                                    f              p                                      )            
where fs is the resonance frequency and fp is the antiresonance frequency.
This parameter is directly related to the piezoelectric properties of the material and to the vibration mode in question.
Typically, for acoustic waves propagating in AlN (aluminum nitride), the electromechanical coupling coefficient is about 7%, enabling filters to be designed with a bandwidth of about 2%.
A filter is produced by arranging a number of resonators, sometimes accompanied by additional passive elements. Typically, two kinds of resonators for filter design can be distinguished, namely resonators connected in parallel with the signal path, and resonators connected in series on the signal path. They are distinguished in electrical terms by the frequency positioning of their resonance and antiresonance frequencies, generally arranged according to the diagram of FIG. 2, which shows the response of the filter (curve C2F).
In particular, the resonance frequency of the resonators connected in series (curve C2s) is aligned to the antiresonance frequency of the resonators connected in parallel (curve C2p).
Outside their distinctive frequencies, acoustic resonators exhibit capacitive behavior, and therefore exhibit an impedance which depends primarily on their equivalent static capacitance. This impedance, related to the capacitive behavior of a resonator outside its distinctive frequencies, is referred to below as the “characteristic impedance Zc”. The design of this capacitor has an effect on the impedance matching of the filter. It can be shown that, when the series and parallel resonators are arranged according to the principle described above, filter matching is achieved if the impedance Zeq of the generator and of the load is equal to the geometric mean of the characteristic impedances of the resonators:Zeq=√{square root over (Zseries·Zseries)}
For example, FIG. 3 shows the topology of a duplexer (an association of two filters for information exchange in frequency duplex mode). Two groups of resonators are identified here, as well as some passive elements.
Whereas the arrangement of the resonators on the “RX” path is differential and intersecting (called a “lattice” arrangement in English), the resonators on the TX path are in what is called a “ladder” arrangement in English.
A few manufacturers currently dominate the market for BAW- or SAW-based filter/duplexers. All of these, without exception, offer sets of components, each component being designed for a specific frequency band or a specific standard (in the case of the duplexers). This is necessitated by the intrinsic properties of the piezoelectric material, and also by the architectural approach followed for the filters, as explained below in the present description.
A proposal to modify the distinctive frequencies of acoustic resonators was published in 2005, thus paving the way for variable filters, as described in the paper by Carpentier, J. F., Tilhac, C., Caruyer, G., and Dumont, F., “A tunable bandpass BAW-filter architecture and its application to WCDMA filter”, 2005 IEEE MTT-S International Microwave Symposium Digest. The principle is based on the addition of capacitance and inductance, in series and in parallel, to each resonator. The authors assert that a variation of 2% in the operating frequency is possible, but this has never been demonstrated in practice.
In 2006 and 2007, however, the same team showed that the central frequency of a filter could be modified by 0.3%, using an active circuit. They concluded that the method could compensate for dispersions due to the production process. It can be assumed that this method is unlikely to lead to true agility in the filter, as described in the paper by Razafimandimby, S., Tilhac, C., Cathelin, A., and Kaiser, A., “An Electronically Tunable Bandpass BAW-Filter for a Zero-IF WCDMA Receiver” (FIG. 4 is taken from this paper), Proceedings of the 32nd European Solid-State Circuits Conference, 2006 (ESSCIRC 2006), and in the paper by Cyrille Tilhac, Andreia Cathelin, Andreas Kaiser, and Didier Belot, “Digital tuning of an analog tunable bandpass BAW-filter at GHz frequency”, 33rd European Solid State Circuits Conference, 2007 (ESSCIRC 2007).
Based on the conclusions of this study, most of the teams working on this topic have searched for solutions for dispensing with the passive elements added to these resonators, by concentrating on means for modifying the propagation speed of acoustic waves.
Thus resonators based on the use of electrostrictive materials have been proposed, as described in the paper by S. Gevorgian, A. Vorobiev, and T. Lewin, “DC field and temperature dependent acoustic resonances in parallel-plate capacitors based on SrTiO3 and Ba0.25Sr0.75TiO3 films: experiment and modeling”, Journal of Applied Physics 99, 124112 (2006). This is because these materials have the property of exhibiting a variation of elastic rigidity as a function of an applied electrical field. They also exhibit an effect which is equivalent to piezoelectricity, but can be intensity modulated, again as a function of an applied electrical field. However, their dielectric properties are also affected by this electrical field, preventing these components from having a constant characteristic impedance. It has been clearly stated that this variation of characteristic impedance makes it impossible to manufacture a filter based on this type of resonator. Moreover, the losses of materials having these properties, namely compounds of the BST (BaxSr1-xTiO3) or PZT (PbxZr1-xTiO3) type, are too high for the production of resonators for practical use.
Another proposed solution is that of producing what are known as composite resonators, based on the stacking of two piezoelectric layers. One of the layers is connected to the user circuit, while the other layer is connected to a tuning circuit, usually a variable capacitor for modifying the conditions at the electrical limits encountered by the bulk acoustic wave as it is propagated in the stack. Thus this structure can be used to provide a frequency-agile resonator which is controllable exclusively by a variable capacitor, as described in the paper by R. Aigner, “Tunable acoustic RF-filters: discussion of requirements and potential physical embodiments”, Proceedings of the 40th European Microwave Conference, p. 787, 2010. In this structure, highly piezoelectric materials such as lithium niobate must be used in order to provide resonators having a electromechanical coupling coefficient which is sufficient for the production of a filter, and frequency agility which is sufficient to cover a number of communication bands. However, the drawback of this structure is that it is extremely complicated, and therefore costly, to produce, since it requires the combination of two piezoelectric layers. On the other hand, this structure provides simple frequency translation of the resonator, but without allowing the independent control of the resonance or antiresonance frequency, which may prove problematic, since not all telecommunication bands have the same width: consequently, a filter cannot simply be frequency translated.
The most promising approach is that which has been followed for several years by a Japanese team led by Ken-Ya Hashimoto of the University of Chiba, amply described in the following papers: Tomoya Komatsu, Ken-ya Hashimoto, Tatsuya Omori, and Masatsune Yamaguchi, “Tunable Radio-Frequency Filters Using Acoustic Wave Resonators and Variable Capacitors”, Japanese Journal of Applied Physics 49 (2010); T. Yasue, T. Komatsu, N. Nakamura, K. Hashimoto, “Wideband tunable love wave filter using electrostatically-actuated MEMS variable capacitors integrated on lithium niobate”, 16th International Solid-State Sensors, Actuators and Microsystems Conference (TRANSDUCERS), 2011; Ken-Ya Hashimoto, S. Tanaka, M. Esashi, “Tunable RF SAW/BAW filters: Dream or reality?”, Joint Conference of the IEEE International Frequency Control and the European Frequency and Time Forum (IFCS), 2011; M. Inaba, K.-Y. Hashimoto, T. Omori, C. Ahn, “A widely tunable filter configuration composed of high Q RF resonators and variable capacitors”, European Microwave Integrated Circuits Conference (EuMIC), 2013, Hideki Hirano, Tetsuya Kimura, Ivoyl P Koutsaroff, Michio Kadota, Ken-ya Hashimoto, Masayoshi Esashi and Shuji Tanaka, “Integration of BST varactors with surface acoustic wave device by film transfer technology for tunable RF filters”, Journal of Micromechanics and Microengineering 2013.
This team is engaged in demonstrating that the first approach mentioned above is in fact feasible, since resonators produced with materials having a very high electromechanical coupling coefficient are available. Indeed, the addition of variable passive elements, in series or in parallel, to an acoustic resonator enables the resonance and antiresonance frequencies to be shifted as shown in FIGS. 5a and 5b, which illustrate the effect of the capacitor in parallel (variation of capacitance Cp from 0 to 12 pF) and that of the capacitor in series (variation of capacitance Cs from ∞ to 12 pF) on the electrical response of a resonator, and more precisely on the modulus of the impedance expressed in Ω, corresponding to the notation mag(Z(1,1)) of FIGS. 5a and 5b. This is because a capacitor in parallel with a resonator creates an increase in the static capacitance of the resonator, causing a displacement of the zero susceptance of the resonator (that is to say the antiresonance) while leaving the pole (that is to say the resonance) unchanged. Conversely, the addition of a capacitor in series with a resonator creates a decrease in its characteristic impedance, and causes a displacement of the zero reactance (that is to say the resonance) while leaving the pole (that is to say the antiresonance) unchanged. The resonance and antiresonance frequencies of the resonator can therefore be shifted substantially within the limits of the range delimited by the resonance and antiresonance frequencies of the original resonator without the adjusting capacitive elements (the curve drawn in solid lines). A very high electromechanical coupling coefficient becomes the necessary condition for very high frequency agility.
The topologies shown in FIGS. 6a and 6b for making use of this principle have been known since 2010, when they were described in the paper by Tomoya Komatsu, Ken-ya Hashimoto, Tatsuya Omori, and Masatsune Yamaguchi, “Tunable Radio-Frequency Filters Using Acoustic Wave Resonators and Variable Capacitors”, Japanese Journal of Applied Physics 49 (2010), and are very similar to those investigated by French teams. The authors of this study have taken the approach of using components making use of surface acoustic waves having the highest possible electromechanical coupling coefficient (over 30% in this case), by selecting lithium niobate substrates from a section not normally used by SAW filter designers, since its electromechanical coupling coefficients are too high to allow filter synthesis in the conventional manner.
Depending on the way in which resonators and capacitors are associated, as shown in the topology of FIG. 6a or FIG. 6b, the authors consider that the type of response shown in FIGS. 7a and 7b, respectively, can be obtained.
The filter of FIG. 7a has a relative bandwidth of approximately 17%, based on the use of a piezoelectric material with k2 equal to 28%. It can be seen that, according to the method indicated in FIG. 6a, the band of the filter is reduced at its lower edge, while being only very slightly increased at its upper edge. In this case, therefore, the aim is to reduce the bandwidth by essentially shifting one edge. This causes the effective central frequency to be modified equally, with a rate of about 3% in this case.
In the case of FIG. 7b, the authors construct a filter with a width of 5%, and make the central frequency vary by 5%. Two filters are therefore seen side by side. Agility, in the true sense of the word, is therefore demonstrated: the method is functional.
However, the application of this solution is extremely limited. In fact, the authors of the study provide two filters with a band of 5%, centered on 4% and 9% of the resonance frequency of the series resonators. In other words, they do not attempt to cover the whole spectrum that might be expected to be usable, in view of the 30% coupling coefficient. A more detailed analysis of these papers shows that, very rapidly, the values of the associated capacitors become such that the characteristic impedance of the filter varies in such a pronounced way that its matching is no longer assured. This can be seen in FIG. 8 (where all the components are assumed to be lossless). The horizontal axis shows a relative frequency, equivalent to a percentage of the natural resonance of the series resonators. Thus the middle filter band is centered on 0.06, that is to say 6% of the natural resonance frequency of the series resonators.
The shifting of the resonances and antiresonances over a large range of variation (at the limit, over the whole range allowed by the coupling coefficient) is based on the use of variable capacitors whose values are either very large or very small relative to the natural capacitance of the resonators. Consequently the matching conditions are no longer met, and the insertion losses of the filter increase. This can be seen, notably, in the curves centered on 0% and 11%.
The mismatching also creates a standing wave ratio (VSWR) at the inputs and outputs of the filter. A generally satisfactory standing wave ratio is equal to 2, corresponding to a reflection coefficient of −10 dB. Many systems tolerate a VSWR of 2.5, or even −7.5 dB. However, filters located on the power transmission path are very strict, and the lowest possible VSWR is desirable (the theoretical minimum is 1). In the filters located at 0% and 11% in FIG. 8, the VSWR is well over 20.
In this context, and in order to gain a clearer understanding of the problems, the present applicants conducted a more detailed study of the example of configuration shown in FIG. 9, which represents a “ladder” filter architecture, shown between the references Num4 and Num5.
On the basis of this architecture, the applicants have attempted to handle a number of frequency bands, namely the TX bands numbered 28, 17, 13, 8 and 5 of the LTE protocol, these bands being shown in FIGS. 10, 11, 12, 13 and 14 respectively. Their simultaneous coverage requires a filter capable of movement over a frequency range from 734 to 960 MHz, produced on the basis of a single resonator: only the series and parallel capacitors change from one band to another. The resonator was designed in an optimal manner (that is to say, for matching the filter to the generator and the load) over band 28 (758-803 MHz):
FIGS. 10a, 11a, 12a, 13a and 14a relate to the transmission response of the filter (S21 or S12);
FIGS. 10b, 11b, 12b, 13b and 14b relate to the reflection response of the filter (S11 or S22), or more precisely:                the curves C10b1, C11b1, C12b1, C13b1 and C14b1 relate to the reflection response S(4,4), expressed in dB, of the filter shown in FIG. 9;        the curves C10b2, C11b2, C12b2, C13b2 and C14b2 relate to the reflection response S(5,5), expressed in dB, of the filter shown in FIG. 9;        
FIGS. 10c, 11c, 12c, 13c and 14c relate to the standing wave ratio (VSWR) calculated at the port Num4;
FIGS. 10d, 11d, 12d, 13d and 14d show:                the curves C10d1, C11d1, C12d1, C13d1 and C14d1 relating to the impedance of a resonator known as a “relaxed” resonator, that is to say one having no variable capacitor, this impedance being determined by its dimensions and the technological characteristics of the piezoelectric layer;        the curves C10d2, C11d2, C12d2, C13d2 and C14d2 relating to the impedance response of the parallel component, composed of the “relaxed” resonator and the associated capacitors;        the curves C10d3, C11d3, C12d3, C13d3 and C14d3 relating to the response of the series component, composed of a resonator identical to that of the parallel component, and associated capacitors which differ from those of the parallel component.        
Points m7 and m9 relate to frequencies of 689 MHz and 923 MHz respectively, and have respective impedances of −6.5 dB and 90 dB.
As anticipated by the prior art, it is found that, when an optimal resonator is defined for band 28, it is possible to achieve, for example, the provision of band 17 (734-746 MHz), whereas the other bands cannot be provided in correct conditions, that is to say with a standing wave ratio of about 2.
The applicants have also conducted a study by selecting an optimal resonator for the center of the range of variation, that is to say centered on 800 MHz (allowing optimal operation in band 13 and three times as much as that calibrated for band 28): the same conclusion is reached. The conclusions would be the same if the study were based on an optimization on the band with the highest frequency.
FIGS. 15a, 15b, 15c and 15d relate, respectively, to the insertion losses, to the matching, to the impedance (without series capacitor and without parallel capacitor) and to the impedance of the pairs of resonant circuits (with variable values of series capacitance and parallel capacitance) for a variable filter centered on 800 MHz. FIG. 15d clearly shows a problem of impedance variation for the different curves, relating to a frequency shift of about 60 MHz around the central frequency of 800 MHz.
The constraint arises not from the operating frequency, but from the frequency range that is to be covered, and the bandwidth of each filter that is to be provided.
An inspection of the impedance curves of the resonators reveals that, when there is a movement in frequency, from the series resonance frequency to the parallel resonance frequency of the “relaxed” resonator, the characteristic impedance of the resonant circuit increases. The resonant circuits of the filter centered on 800 MHz exhibit a characteristic impedance of about 50 ohms, while they vary by about 10 ohms and about 200 ohms, respectively, at −60 MHz and +60 MHz. This dispersion of characteristic impedance is the main cause of the limitations of this approach.
For this reason, and in this context, the applicants propose a new solution for stabilizing the characteristic impedance of a resonant circuit comprising a resonator at a chosen value and making it possible, notably, to produce filters with an adjustable central frequency and an equally adjustable band, while ensuring their impedance matching.
The following description explains the inventive reasoning followed by the applicants which has enabled them to develop the solution according to the present invention.
Starting from the aforementioned problems of the prior art solutions, the applicants have investigated a method counter to the prior art, and, rather than causing the capacitances associated with a fixed resonator to vary, the applicants have studied the operation of a BAW filter constructed on the basis of a resonator (single layer) with a variable surface (and therefore a variable static capacitance), and with fixed associated capacitors, even though the variation of the static capacitance of a piezoelectric resonator is not a parameter that is directly imposed. In fact, the surface or thickness of a resonator is determined by the technology, being a matter of the physical dimensions of the component. This option is therefore available for the electrostrictive resonators mentioned in the prior art, although these are accompanied by other constraining effects.
Thus, on the basis of the topology shown in FIG. 9, and with the associated capacitors fixed at the values corresponding to the optimal values for band 13, the applicants have studied the behavior while changing only the surface of the resonator. In this case, they obtained the results shown in FIGS. 16a, 16b, 16c and 16d for resonators with a side measurement varying from 50 μm to 300 μm. FIGS. 16a, 16b, 16c and 16d relate, respectively, to the insertion losses, to the matching, to the impedance (without series capacitor and without parallel capacitor) and to the impedance of the pairs of resonant circuits (with fixed values of series capacitance and parallel capacitance):
FIG. 16a clearly shows that the insertion losses are maintained;
FIG. 16b clearly shows that the matching is maintained;
FIG. 16c clearly shows the variation of the characteristic impedance as a function of the surface;
FIG. 16d clearly shows that the characteristic impedance of the pairs of resonant circuits is maintained.
The smallest “relaxed” resonator is located at a lower frequency, while the largest “relaxed” resonator is located at a higher frequency. It can be seen that the response of the device is such as to produce a filter of about 20 MHz (that is to say, 2.5% of bandwidth at 800 MHz), matched from 725 MHz to 875 MHz, that is to say more than 18% around 800 MHz. On the other hand, the plot of the impedance of the series and parallel resonators shows that they all have the same characteristic impedance, located around 50 ohms. FIG. 16d is remarkable in that it demonstrates this stability of impedance.
Finally, it is also very interesting to note that the relative positioning of the distinctive frequencies of the series and parallel resonators is preserved, regardless of the chosen surface of the “relaxed” resonator.
Therefore the filter constructed in this way has the property of having a fixed bandwidth and a variable central frequency.
The applicants have thus demonstrated that it becomes possible to use a simple elementary component (a simple layer of material) having a high coupling coefficient (for example, with materials such as lithium niobate (LiNbO3, or LNO) or potassium niobate (KNbO3)) and a variable impedance to achieve what was proposed in the prior art using composite resonators as described in the paper by A. Reinhardt, E. Defaÿ, F. Perruchot, C. Billard, “Tunable composite piezoelectric resonators: a possible ‘Holy Grail’ of RF filters?”, Proceedings of the International Microwave Symposium, 2012.
Furthermore, this new “variable impedance resonator” can also be associated with a variable reactive component in series and another in parallel. In fact, this association makes it possible to have complete freedom as regards the frequency positioning of the resonance, the antiresonance, and the impedance of the assembly. Therefore it becomes possible to construct filters with variable bandwidth and a variable central frequency.
Thus, on the basis of the common filtering topology at any band located between 700 MHz and 850 MHz, for example the TX bands 28, 17, 13 and 5, shown in FIG. 17, the applicants have demonstrated that it is, notably, possible to construct, for example, filters suitable for the TX bands numbered 28, 17, 13 and 5, and meeting the specifications for a possible production of 4 duplexers in one, by adjusting the surface of each resonator, as well as the associated capacitors in series and in parallel.
More precisely, the table below shows all the values of surface, series capacitors and parallel capacitors used in the topology shown in FIG. 17, to provide the filtering functions shown in FIG. 18, configured for the TX bands 28, 17, 13 and 5.
Band 28Band 13P = 70 × 70 μmP2 = 80 × 80 μmP = 250 × 250 μmP2 = 250 × 250 μmCs(P) = ∞Cs(P2) = ∞Cs(P) = 15.2 pFCs(P2) = 15.5 pFCp(P) = 6.7 pFCp(P2) = 7.5 pFCp(P) = 18 pFCp(P2) = 18 pFS = 105 × 105 μmS2 = 55 × 55 μmS = 200 × 200 μmS2 = 150 × 150 μmCs(S) = 11.8 pFCs(S2) = 4.2 pFCs(S) = 4.8 pFCs(S2) = 2.6 pFCp(S) = 6.8 pFCp(S2) = 1.4 pFCp(S) = 10.3 pFCp(S2) = 5.8 pFBand 17Band 5P = 100 × 100 μmP2 = 100 × 100 μmP = 300 × 300 μmP2 = 300 × 300 μmCs(P) = ∞Cs(P2) = ∞Cs(P) = 11.1 pFCs(P2) = 11.2 pFCp(P) = 12.1 pFCp(P2) = 14 pFCp(P) = 11.9 pFCp(P2) = 12.1 pFS = 70 × 70 μmS2 = 50 × 50 μmS = 180 × 180 μmS2 = 125 × 125 μmCs(S) = 5.7 pFCs(S2) = 2.8 pFCs(S) = 3.7 pFCs(S2) = 1.7 pFCp(S) = 5.2 pFCp(S2) = 2.4 pFCp(S) = 2.1 pFCp(S2) = 1.1 pF
The applicants have thus been able to demonstrate that the same stacking method and the same topology can be used to construct, notably, the RX filters of the same bands 28, 17, 13 and 5.
The approach described above proves that, with the present invention, it becomes possible to design, notably, a reconfigurable filter covering any frequency band located between 700 and 900 MHz, that is to say a frequency coverage of 25%.